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AND8016/D Design of Power Factor Correction Circuit Using GreenlineTM Compact Power Factor Controller MC33260
Prepared by Ming Hian Chew ON Semiconductor Analog Applications Engineering
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APPLICATION NOTE
of external components, the MC33260 can control the follower boost operation that is an innovative mode allowing a drastic size reduction of both the inductor and the power switch. Ultimately, the solution system cost is significantly lowered. Also able to function in a traditional way (constant output voltage regulation level), any intermediary solutions can be easily implemented. This flexibility makes it ideal to optimally cope with a wide range of applications. This application note will discuss on the design of power factor correction circuit with MC33260 with traditional boost constant output voltage regulation level operation and follower boost variable output voltage regulation level operation. For derivation of the design equations related to the IC please refer to MC33260 data sheet.
R6 R7 D5
Introduction The MC33260 is an active power factor controller that functions as a boost pre-converter which, meeting international standard requirement in electronic ballast and off-line power supply application. MC33260 is designed to drive a free running frequency discontinuous mode, it can also be synchronized and in any case, it features very effective protections that ensure a safe and reliable operation. This circuit is also optimized to offer extremely compact and cost effective PFC solutions. It does not entail the need of auxiliary winding for zero current detection hence a simple coil can be used instead of a transformer if the MC33260 Vcc is drawn from the load (please refer to page 19 of the data sheet). While it requires a minimum number
D1 D2 D3 D4 C1
D7
+ C4
L1
D5 1 2 C2 R2 C3 R1 R3 C6 R4 MC33260 3 4 6 5 8 R5 7 Q1 + C5
Figure 1. Application Schematic of MC33260
PFC Techniques Many PFC techniques have been proposed, boost topology, which can operate in continuous and discontinuous mode, is the most popular. Typically, continuous mode is more favorable for high power application for having lower peak current. On the other hand, for less than 500 W application, discontinuous mode offers smaller inductor size, minimal parts count and lowest
(c) Semiconductor Components Industries, LLC, 2002
cost. This paper will discuss design of PFC with MC33260, which operates in critical conduction mode. Discontinuous Conduction Mode Operation Critical conduction mode operation presents two major advantages in PFC application. For critical conduction mode, the inductor current must fall to zero before start the next cycle. This operation results in higher efficiency and
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Publication Order Number: AND8016/D
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eliminates boost rectifier reverse recovery loss as MOSFET cannot turn-on until the inductor current reaches zero. Secondly, since there are no dead-time gaps between cycles, the ac line current is continuous thus limiting the peak switch to twice the average input current. The converter works right on critical conduction mode, which results in variable frequency operation. Inductor Waveform
V + di L dt (1) where I inpk + 2I inrms (5)
Input power of the PFC circuit, Pin can be expressed in following equation, by substituting equation (3) and (5).
P +V I + in inrms inrms Po + V oI o + P in V inpk 2
@
I
inpk 2
+
V I inpk inpk 2
(6)
The output power, Po is given by:
(7)
Equation (1) is the center of the operation of PFC boost converter where V=Vin(t), the instantaneous voltage across the inductor. Assuming the inductance and the on-time over each line half-cycle are constant, di is actually the peak current, ILpk, this is because the inductor always begins charging at zero current.
Vinpk ILpk Vin(t) IL(t)
PFC circuit efficiency is needed in the design equation, for low line operation, it is typically set at 92% while 95% for high line operation. Substituting equation (6) into equation (7),
Po + P + in V I inpk inpk 2 (8)
Express the above equation in term of Iinpk,
I inpk + 2Po 2 Po + V V inpk inrms (9)
Iinpk
Iin(t)
The average input current is equal to average inductor current, IL(avg),
I L(avg) +I in (10)
MOSFET
ON OFF
It has been understood that peak inductor current, ILpk is exactly twice the average inductor current, IL(avg) for critical conduction operation.
I Figure 2. Inductor Waveform Lpk + 2I L(avg) + 2 2 Po V inrms (11)
Design Criteria The basic design specification concerns the following: * Mains Voltage Range: Vac(LL) - Vac(HL) * Regulated DC Output Voltage: Vo * Rated Output Power: Po * Expected Efficiency, h PFC Power Section Design Instantaneous Input Voltage, Vin(t) Peak Input Voltage, Vinpk Both Vin(t) and Vinpk are related by below equation
V (t) + V sin(t) in inpk where V inpk + 2V inrms (2) (3)
Since ILpk is maximum at minimum required ac line voltage, Vac(LL),
I Lpk + 2 2 Po V ac(LL) (12)
Switching Time In theory, the on-time, t(on) is constant. In practice, t(on) tends to increase at the ac line zero crossings due to the charge on output capacitor Cout. Let Vac = Vac(LL) for initial t(on) and t(off) calculations. On-time By solving inductor equation (1), on-time required to charge the inductor to the correct peak current is:
t (on) +I L P Lpk Vinpk (13)
Instantaneous Input Current, Iin(t) Peak Input Current, Iinpk, Both Iin(t) and Iinpk are related by below equation
I (t) + I sin(t), in inpk (4)
Substituting equation (3) and (12) into equation (13), results in:
t (on) + 2 2 Po V ac(LL)
@
L
P
2 Vac(LL)
+
2Po L P V 2ac(LL)
(14)
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Off-time The instantaneous switch off-time varies with the line and load conditions, as well as with the instantaneous line voltage. Off-time is analyzed by solving equation (1) for the inductor discharging where the voltage across the inductor is Vo minus Vin.
t (off) + L Lpk P Vo * V sin(t) inpk I (15)
The exact inductor value can be determined by solving equation (21) by substituting equation (19) and (20) at the selected minimum operating frequency.
t total +t (on)max )t (off)max (21)
Equation (21) becomes
t total + 2 P oL V o P V 2 o*V V ac(LL) ac(LL) 2 (22)
Multiplying nominator Vinpksinw(t) results in:
I
and
denominator
with
By rearranging in term of Lp,
t
V
L Lpk P V sin(t) inpk t + + (off) Vo * V sin(t) inpk V sin(t) inpk
t
(on) *1
(16)
Lp +
total
2 o*V V ac(LL) ac(LL) 2 2 Vo Po
(23)
Vo 2 Vinpk sin()
Equation (23) can be rewritten by substituting rearranged equation (12) in term of 2Po.
2 Lp + t
V
where wt = q The off-time, t(off) is greatest at the peak of the ac line voltage and approaches zero at the ac line zero crossings. Theta (q) represents the angle of the ac line voltage. The off-time is at a minimum at ac line crossings. This equation is used to calculate t(off) as Theta approaches zero.
I t (off)min + L Lpk P , + 0 Vo (17)
total
o*V V ac(LL) ac(LL) 2 Vo I Lpk
(24)
Let the switching cycle t = 40ms for universal input (85 to 265 Vac) operation and 20 ms for fixed input (92 to 138 Vac, or 184 to 276 Vac) operation. Inductor Design Summary The required energy storage of the boost inductor is:
2 W + 1L I L 2 P Lpk (25)
Switching Frequency
f+ t (on) 1 )t (off) (18)
The number of turns required for a selected core size and material is:
LI 10 6 P Lpk N+ P B maxA e (26)
Switching frequency changes with the steady state line and load operating conditions along with the instantaneous input line voltage. Typically, the PFC converter is designed to operate above the audible range after accommodating all circuit and component tolerances. 25 kHz is a good first approximation. Higher frequency operation that can significantly reduce the inductor size without negatively impacting efficiency or cost should also be evaluated. The minimum switching frequency occurs at the peak of the ac line voltage. As the ac line voltage traverses from peak to zero, t(off) approaches zero producing an increase in switching frequency. Inductor Value Maximum on-time needs to be programmed into the PFC controller timing circuit. Both t(on)max and t(off)max will be individually calculated and added together to obtain the maximum conversion period, ttotal. This is required to obtain the inductor value.
t (on)max + 2Po L P V 2ac(LL) L Lpk P , () [ 90 Vo * V inpk I (19)
where Bmax is in Teslas and Ae is in square millimeters (mm2) The required air gap to achieve the correct inductance and storage is expressed by:
l gap + * 2 4p10 7 N p A e mm L P (27)
Design of Auxiliary Winding MC33260 does not entail an auxiliary winding for zero current detection. Hence if DC voltage can be tapped from the SMPS or electronic ballast connected to the output of PFC, this step can be skipped. Then an inductor is what it needs. The auxiliary winding exhibits a low frequency ripple (100-120 Hz). The Vcc capacitor must be large enough (about 47 mF) to minimize voltage variations. As a rule of thumb, you can use the below equation to estimate the auxiliary turn number:
Naux + Np @ Vaux Np @ Vaux + Vo * Vac(HL) VL (28)
t
(off)max
+
(20)
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The MC33260 VCC maximum voltage being 16 V, one must add a resistor (in the range of 22 W) and a 15 V zener to protect the circuit against excessive voltages. Vaux should be chosen above the Under-Voltage Lockout threshold (10 V) and below the zener voltage. Selection of Output Capacitor The choice of output capacitance value is dictated by the required hold-up time, thold or the acceptable output ripple voltage, Vorip for a given application. As a rule of thumb, can start with 1 mF/watt. Selection of Semiconductors Maximum currents and voltages must first be determined for over all operating conditions to select the MOSFET and boost rectifier. As a rule of thumb, derate all semiconductors to about 75-80% of their maximum ratings. This implying the need of devices with at least 500 V breakdown voltage. Bipolar transistors are an acceptable alternative to MOSFET if the switching frequency is maintained fairly low. High voltage diodes with recovery times of 200 ns, or less should be used for the boost rectifier. One series of the popular devices is the MURXXX Ultrafast Rectifier Series from ON Semiconductor. Maximum power MOSFET conduction losses.
P [1 (on)max 6 R ds(on) I 1.2 2 1* Lpk V ac(LL) (29) Vo
Overcurrent protection resistor, ROCP can be determined with below equation:
R + OCP R I Lpk CS I OCP (33)
Current Limiting With Boost Topology Power Factor Correction Circuit Unlike buck and flyback circuits, because there is no series switch between input and output in the boost topology, high current occurring with the start-up inrush current surge charging the bulk capacitor and fault load conditions cannot be limited or controlled without additional circuitry. The MC33260 Zero Current Detection uses the current sensing information to prevent any power switch turn on as long as some current flows through the inductor. Then, during start-up, the power MOSFET is not allowed to turn on while in-rush current flows. Then there is no risk to have the power switch destroyed at start-up because of the in-rush current. In the same way, in an overload case, the power MOSFET is kept off as long as there is a direct output capacitor charge current, i.e., when the input voltage is higher than the output voltage. Consequently, overload working is fully safe for the power MOSFET. This is one of the major advantages compared to MC33262 and competition. Current Limiting for Start-up Inrush Initially Vo is zero, when the converter is turned on, the bulk capacitor will charge resonantly to twice Vin. The voltage can be as high as 750 V if Vin happens to be at the peak high-line 265 V condition (375 V). The peak resonant charging current through the inductor will be many times greater than normal full load current. the inductor must be designed to be much larger and more expensive to avoid saturation. The boost shunt switch cannot do anything to prevent this and could be worse if turned on during start-up. The inrush current and voltage overshoot during the start-up phase is intolerable. A fuse is not suitable, as it will blow each time the supply is turned on. There are several methods that may be used to solve the start-up problem: 1. Start-up Bypass Rectifier This is implemented by adding an additional rectifier bypassing the boost inductor. The bypass rectifier will divert the start-up inrush current away from the boost inductor as shown in Figure 3. The bulk capacitor charges through Dbypass to the peak AC line voltage without resonant overshoot and without excessive inductor current. Dbypass is
Designing the Oscillator Circuit For traditional boost operation, CT is chosen with below equation:
Cw T 2 K osc L P 2 V ac(LL) P in 2 Ro V2 o *C int (30)
Design of Regulation and Overvoltage Protection Circuit The output voltage regulation level can be adjusted by Ro,
Vo Ro [ 200 mA (31)
Designing the Current Sense Circuit The inductor current is converted into a voltage by inserting a ground referenced resistor, RCS in series with the input diode bridge. Therefore a negative voltage proportional to the inductor current is built. The current sense resistor losses, PRcs:
P +1 Rcs 6 R CS 2 I Lpk (32)
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reverse-biased under normal operating conditions. If load overcurrent pulls down Vo, Dbypass conducts, but this is probably preferable to having the high current flowing through boost inductor.
Dbypass
* Regulated DC Output Voltage: Vo = 400 Vdc * Rated Output Power: Po = 80 W * Expected Efficiency, h > 90%
A. The input power, Pin is given by
P P + o + 80 + 86.96 W in 0.92
PFC IC VAC
+
VOUT
B. Input diode current is maximum at Vinrms = Vac(LL)
I inpk + 2 Po 2 + 0.92 V ac(LL) 80 + 1.447 A 85
Figure 3. Rectifier bypass of start-up inrush current
C. Inductor design 1. Inductor peak current:
I Lpk + 2I inpk +2 1.447 + 2.894 A
2. External Inrush Current Limiting Circuit For low power system, a thermistor in series with the pre-converter input will limit the inrush current. Concern is the thermistor may not respond fast enough to provide protection after a line dropout of a few cycles. A series input resistor shunted by a Triac or SCR is a more efficient approach. A control circuit is necessary. This method can function on a cycle-by-cycle basis for protection after a dropout. Load Overcurrent Limiting If an overcurrent condition occurs and exceeds the boost converter power limit established by the control circuit, Vo will eventually be dragged down below the peak value of the AC line voltage. If this happens, current will rise rapidly and without limit through the series inductor and rectifier. This may result in saturation of the inductor and components will fail. The control circuit holds off the shunt switch, since the current limit function is activated. It cannot help to turn the switch ON - the inductor current will rise even more rapidly and switch failure will occur. Typically, a power factor correction circuit is connected to another systems like switched mode power supply or electronic ballast. These downstream converters typically will have current limiting capability, eliminating concern about load faults. However, a downstream converter or the bulk capacitor might fail. Hence there is a possibility of a short circuit at the load. If it is considered necessary to limit the current to a safe value in the event of a downstream fault, some means external to the boost converter must be provided. Design Example I - Traditional Boost Constant Output Voltage Regulation Level Operation Power Factor Correction The basic design specification concerns the following: * Mains Voltage Range: Vac(LL) - Vac(HL) = 85 - 265 Vac
2. Inductor value:
2 Lp + 2 + t
V
total
o*V V ac(LL) ac(LL) 2 Vo I Lpk
40
10 *6 400
400 2
* 85 85
2.894
+ 1.162 mH
Let the switching cycle t = 40 ms for universal input (85 to 265 Vac) operation. 3. The number of turns required for a selected core size and material is:
LI 10 6 P Lpk N+ + 1.162 P B maxA e 10 *3 0.3 2.894 60 10 *6
+ 186.8 turns [ 187 turns
Using EPCOS E 30/15/7, Bmax =0.3 T and Ae = 60 mm2. 4. The required air gap to achieve the correct inductance and storage is:
l gap + 4p10 *7 2 N p Ae L P 10 *7 187 2 1.162 + 2.269 mm
+ 4p
60 *3 10
10 *6
5. Design of Auxiliary Winding
N aux + Vaux N P Vo * V ac(HL) + 14 187 (400 * 265)
+ 19.4 turns [ 20 turns
Round up to 20 turns to make sure enough voltage at the auxiliary winding.
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D. To determine the output capacitor As rule of thumb, for 80 W output, start with 100 mF, 450 V capacitor. E. Calculation of MOSFET conduction losses A 8A, 500V MOSFET, MTP8N50E is chosen. The on resistance, Rds(on) [ 1.75 W@100C. Therefore, maximum power MOSFET conduction losses is:
P [1 (on)max 6 +1 6 R ds(on) 1.2 2 I Lpk 1 * V ac(LL) Vo
445 415 385 355 325 295 265 235 205 175 145 115 85 Vo/(V)
1.75
2.894 2 1 * 1.2 85 + 1.82 W 400
Full Load Half Load Vacpeak 85 100 115 1130 145 160 175 190 205 220 235 250 265 280 Vac (V)
F. Design of regulation and overvoltage protection circuit The output voltage regulation level can be adjusted by Ro,
Vo Ro [ + 400 + 2 M 200 A 200 A
Figure 4. Theoretical Vo versus Vac with CT = 10nF
Use two 1 MW resistors in series. G. Designing the oscillator circuit For traditional boost operation, CT is chosen with below equation:
Cw T 2 2 K osc L P V 2ac(LL) P in R2 o V2 o *C + int 400 2 * 15pF + 7.16nF
H. Design of the current sense circuit Choose Rcs = 0.68 W 1. So the current sense resistor losses, PRcs:
P +1 Rcs 6 R CS 2 I Lpk + 1 6 1 2 2.894 + 0.949 W
Therefore the power rating of RCS is chosen to be 2 W. 2. Overcurrent protection resistor, ROCP can be determined with below equation:
R + OCP R I Lpk CS + 0.68 2.894 + 9600 I 205 A OCP
6400
1.162mH 86.96 85 2 2M 2
Use 10 nF capacitor.
Use 10000 W resistor. This provide current limit at 3.01 A versus calculated value of ILpk = 2.894 A.
80 W, Universal Input, Traditional Boost Constant Output Voltage Level Regulation Operation Power Factor Correction Circuit Part List
Index C1 C2 C3 C4 C5 C6 R1 R2 R3 R4 R5 Value 0.63 mF@600 V 680 nF 10 nF 100 mF@50 V 100mF@450V 1 nF@50 V 0.68 W@2 W 10 KW@0.25 W 1 MW@0.25 W 1 MW@0.25 W 10 W@0.25 W Comment Filtering Capacitor Pin 2 Vcontrol Capacitor Pin 3 Oscillator Capacitor Aux Capacitor, E-Cap Output Capacitor, E-Cap Feedback Filtering Capacitor Current Sense Resistor OCP Sensing Resistor Feedback Resistor Feedback Resistor Gate Resistor Index R6 R7 R8 D1 D2 D3 D4 D5 D6 D7 Q1 Value 22 W@0.25 W 100 KW@2 W 1N5406 1N5406 1N5406 1N5406 1N4937 MUR460 1N5245 MTP8N50E 1.162 mH Comment Aux Winding Resistor Start-up Resistor Input Diode Input Diode Input Diode Input Diode Aux Winding Diode Boost Diode Aux 15 V Zener Diode Power MOSFET Inductor
* E 30/15/7, N67 Material from EPCOS Primary - 187 turns of # 23 AWG, Secondary - 19 turns of # 23 AWG. Gap length 2.269mm total for a primary inductance LP of 1.162mH.
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D1 D2 D3 D4 C1 D7 + C4 D5 1 2 C2 R2 C3 R1 R3 C6 R4 MC33260 3 4 6 5 8 R5 7 Q1 + C5 R6 R7 L1 D5
Figure 5. 80 W Universal Input, Traditional Boost Constant Output Voltage Regulation Level Operation Power Factor Correction Circuit Design Table for Universal Input, Traditional Boost Constant Output Voltage Regulation Level Operation Power Factor Correction
Po LP Co RCS ROCP Cin CT Q Dout Din 25 3.720 33 2 10000 0.22 10 MTP4N50E MUR160 1N4007 50 1.860 68 1 10000 0.63 10 75 1.240 100 0.68 10000 0.63 10 100 0.930 100 0.5 9100 1.0 10 MTP8N50E MUR460 1N5406 125 0.744 150 0.39 9100 1.0 10 150 0.620 150 0.33 9100 1.0 10 200 0.465 220 0.25 9100 1.0 10 (Watts) (mH) (mF) W W (mF) (nF)
MTW14N50E
Design Example II - Follower Boost Variable Output Voltage Regulation Level Operation Power Factor Correction The basic design specification concerns the following: * Mains Voltage Range: Vac(LL) - Vac(HL) = 85 - 265 Vac * Maximum Regulated DC Output Voltage: Vo = 400 Vdc * Minimum Regulated DC Output Voltage: Vomin = 140 Vdc * Rated Output Power: Po = 80 W * Expected Efficiency, h > 90% A. The input power, Pin is given by
P P + o + 80 + 86.96 W in 0.92
B. Input diode current is maximum at Vinrms = Vac(LL)
I inpk + V ac(LL) 2 Po + 2 0.92 80 + 1.447 A 85
C. Inductor design 1. Inductor peak current:
I Lpk + 2I inpk +2 1.447 + 2.894 A
2. Inductor value, for follower boost operation, Vo = Vomin:
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V
2 Lp +
t
total
omin * V ac(LL) 2
V I omin Lpk 2 40 10 *6 140
140 2
F. Design of regulation and overvoltage protection circuit The output voltage regulation level can be adjusted by Ro,
Vo Ro [ + 400 + 2 M 200 A 200 A
+
* 85 85
2.894
+ 0.235 H
Use two 1MW resistors in series. G. Designing the Oscillator Circuit For follower boost operation, CT is chosen with below equation:
Cw T 2 2 K osc L P V 2ac(LL) P in R2 o V2 o *C + int 140 2 * 15pF + 162 pF
Let the switching cycle t = 40 ms for universal input (85 to 265 Vac) operation. 3. The number of turns required for a selected core size and material is:
LI P Lpk N+ + P B maxA e 0.235 10 *3 2.894 0.3 32.1 10 6 + 70.6 turns [ 71 turns 10 6
6400
0.234mH 86.96 85 2 2 M 2
Using EPCOS E 20/10/6, N67 material, Bmax =0.3 T and Ae = 32.1 mm2. 4. The required air gap to achieve the correct inductance and storage is:
l gap + 4p10 *7 2 N p Ae L P 10 *7 71 2 0.235 + 0.856 mm
Use 150 pF capacitor.
445 415 385 355 325 295 265 235 205 175 145 115 85 Vo/(V)
+ 4p
32.1 *3 10
10 *6
5. Design of Auxiliary Winding
N aux + Vaux N P Vo * V ac(HL) + 14 71 (400 * 265)
Full Load Half Load Vacpeak 85 100 115 1130 145 160 175 190 205 220 235 250 265 280 Vac (V)
+ 7.4 turns [ 8 turns
Figure 6. Theoretical Vo versus Vac with CT = 150pF
Round up to 8 turns to make sure enough voltage at the auxiliary winding. D. To determine the output capacitor As rule of thumb, for 80 W output, start with 100 mF, 450 V capacitor. E. Calculation of MOSFET conduction losses A 4A, 500 V MOSFET, MTP4N50E is chosen. The on resistance, Rds(on) [ 1.75 W@100C. Therefore, maximum power MOSFET conduction losses is:
P [1 (on)max 6 +1 6 1.2 V 2 ac(LL) R I Lpk 1 * ds(on) V omin 2 1 * 1.2 85 + 0.66 W 1.75 2.894 140
H. Design of the Current Sense Circuit Choose Rcs = 0.68 W 1. So the current sense resistor losses, PRcs:
P +1 Rcs 6 1 + 6 R CS 0.68 2 I Lpk 2.894 2 + 0.949 W
2. Overcurrent protection resistor, ROCP can be determined with below equation:
R + OCP R I Lpk CS + 0.68 2.894 + 9600 I 205 A OCP
Use 10000 W resistor. This provide current limit at 3.01 A versus calculated value of ILpk = 2.894 A.
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80 W, Universal Input, Follower Boost Variable Output Voltage Regulation Level Operation Power Factor Correction Circuit Part List
Index C1 C2 C3 C4 C5 C6 R1 R2 R3 R4 R5 Value 0.63 mF@600 V 680 nF 150 pF 100 mF@50 V 100 mF@450 V 1 nF@50 V 0.68 W@2 W 10 KW@0.25 W 1 MW@0.25 W 1 MW@0.25 W 10 W@0.25 W Comment Filtering Capacitor Pin 2 Vcontrol Capacitor Pin 3 Oscillator Capacitor Aux Capacitor, E-Cap Output Capacitor, E-Cap Feedback Filtering Capacitor Current Sense Resistor OCP Sensing Resistor Feedback Resistor Feedback Resistor Gate Resistor Index R6 R7 D1 D2 D3 D4 D5 D6 D7 Q1 L1* Value 22 W@0.25 W 100 KW@2 W 1N5406 1N5406 1N5406 1N5406 1N4937 MUR460 1N5245 MTP4N50E 0.235 mH Comment Aux Winding Resistor Start-up Resistor Input Diode Input Diode Input Diode Input Diode Aux Winding Diode Boost Diode Aux 15 V Zener Diode Power MOSFET Inductor
* E 20/10/6, N67 Material from EPCOS Primary - 71 turns of # 23 AWG, Secondary - 8 turns of # 23 AWG. Gap length 0.865 mm total for a primary inductance LP of 0.235 mH.
D1 D2
D3 D4 C1
R6 R7 +
D5
L1 C4 D5 + C5
D7
1 2 C2 R2 C3 R1 R3 C6 MC33260 3 4
8 R5 7 6 5 Q1
R4
Figure 7. 80 W Universal Input, Follower Boost Variable Output Voltage Regulation Level Operation Power Factor Correction Circuit
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Design Table for Universal Input, Follower Boost Variable Output Voltage Regulation Level Operation Power Factor Correction
Po LP Co RCS ROCP Cin CT Q Dout Din 25 0.752 33 2 10000 0.22 0.162 50 376 68 1 10000 0.63 0.162 75 0.251 100 0.68 10000 0.63 0.162 100 0.188 100 0.5 9100 1.0 0.162 125 0.150 150 0.39 9100 1.0 0.162 150 0.102 150 0.33 9100 1.0 0.162 MTP8N50E MUR460 1N5406 1N5406 200 0.094 220 0.25 9100 1.0 0.162 (Watts) (mH) (mF) W W (mF) (nF)
MTD2N50E MUR160 1N4007
MTP4N50E
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Notes
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